The prior art describes a variety of methods for converting a temperature sensitive resistance to a voltage. Historically most methods are based on a bridge technique where a device with a known temperature versus resistance characteristic is balanced by a precision resistor, often implemented as a decade resistor box, which would be set by hand or servo to balance the bridge [see e.g. Sensor Technology Handbook, ed. J. S. Wilson, pp. 556-557, Temperature Measurement, Ed. B. G. Liptak, 1993, pp. 79, 92]. FIG. 1 shows an example of a Wheatstone bridge circuit using a decade resistor Rd to balance the thermistor sensor resistance. In the case of FIG. 1, a zero voltage output from the amplifier U1 indicates a balanced bridge. Reading the decade resistor provides the thermistor resistance and the temperature can thus be read from the thermistor temperature versus resistance curve. The two resistors R1 and Rd in the top of the bridge must be matched, but the absolute resistance value of R1 and Rd is not critical to the accuracy of the temperature measurement. VREF, R2 and RT denote a stable reference voltage, a fixed resistance and a thermistor, respectively. This circuit allows a high gain amplifier to be used without concern about amplifier nonlinearity, as only the zero voltage output is used. Driving the amplifier output to zero volts in conjunction with available resistors can sometimes provide sufficient accuracy, noise immunity, and independence from amplifier nonlinearity. However, this prior art approach entails a circuit that is bulky, complex, and slow as it requires the switching of a decade resistor box. Amplifier offset voltages and drift are still a concern, requiring a calibration routine to measure the amplifier offset and also requiring the use of stabilized amplifiers to reduce the drift.
Newer bridge techniques use stabilized amplifiers to provide zero drift and precision bridge resistors instead of a variable tuning resistor. FIG. 2 shows an example of such a circuit. In FIG. 2 U1 denotes the amplifier, R1, R2, R3, RT represent the balanced bridge resistors and Vref designates the reference voltage source. The amplifier has high linearity and amplifies the difference between the thermistor and the reference resistor rather than adjusting a resistor bridge. Even though many thermistors have a large resistance change with temperature, very high (over 40 dB) amplifier gains are required to achieve amplifier outputs on the order of even a few volts over a 0 to 50° C. range. Also, high gain amplifiers create greater sensitivity to amplifier offset and drift. The circuit shown in FIG. 2 also requires a known reference voltage as the absolute value of the output voltage is now of significance. Also, every bridge resistor must be of high precision. [see Sensor Technology Handbook, ed. J. S. Wilson, pp. 556-557, Temperature Measurement, Ed. B. G. Liptak, 1993, pp. 79, 92 and Transducer Interfacing Handbook, Ed. D. Sheingold, 1980, p. 149]. See also U.S. Pat. Nos. 5,537,049; 5,066,140; 4,648,270; 4,161,880; and 3,942,123.
For the prior art circuit shown in FIG. 3 a direct connection to an analog-to-digital converter (ADC), without an analog amplifier, can be used when wire resistance to the sensor is insignificant compared to the sensor resistance. Modern ADCs can be configured as shown in FIG. 3 so that the ADC can measure both the sensor and the reference voltage. [Sensor Technology Handbook, ed. J. S. Wilson, pg. 42]. This eliminates reference accuracy and drift issues but requires that resistors of known resistance, or at least precisely matched resistors, be used in the bridge. Also, mounting the ADC remote from the sensor increases the risk of noise ingress on the bridge voltage and creates an offset in the sensor reading due to the resistance of the wire to the thermistor
If the measurement circuit is remote from the sensor then wire resistance will add to the sensor resistance unless some kind of compensation technique is implemented. Typically sensor wires are very thin to minimize thermal conductivity and heat loss, but this thinness increases electrical resistance in the connecting wires and can thus contribute to measurement errors. FIGS. 4A, 4B, and 4C show two-wire, three-wire, and four-wire connection methods, respectively [see Temperature Measurement, Ed. B. G. Liptak, 1993, pp. 79-81, Transducer Interfacing Handbook, Ed. D. Sheingold, 1980, p. 137, Sensor Technology Handbook, ed. J. S. Wilson, pp. 556-557]. When high resistance thermistors are used, such as those of 10 k ohms and above, this problem can be less significant or even negligible, ultimately depending on the accuracy required. In any case, several methods can be used to compensate for wire resistance. FIG. 4A shows a circuit where the wire resistance (RW) to and from the sensor (RT) adds to the total circuit resistance (R1) and so creates an offset in the temperature conversion. FIG. 4B shows a three-wire technique to compensate for the wire resistance. In FIG. 4B the bridge switches to a loop with the lower two wires and a known resistance being used to calculate the loop resistance. Because all the wires are the same diameter and length, this allows the resistive offset to be computed so that the sensor resistance can be ascertained. In this case the single wire from the sensor (RT) would attach to the bottom of the bridge and the other two wires to a common node would attach to the upper arm of the bridge and to the measurement circuit, respectively. The added resistance of each wire appears equally in the upper and lower bridge paths and so cancels. Because no current flows through the wire to the measurement circuit the three wire configuration can compensate for the sensor wire lengths. FIG. 4C shows a four-wire technique used to mitigate the added resistance of the wire length. In FIG. 4C current to the sensor is sent through the outer pair of wires, while the voltage across the sensor (RT) is read from the inner pair of wires. Because essentially no current flows in the sensor wires the voltage reading across the thermistor indicates the true voltage drop.
Another aspect of prior art concerns the nonlinear resistance versus temperature curve of thermistors. FIG. 5 shows a curve for a typical, as used in the prior art, 10 k ohm thermistor. The thermistor resistance characteristic is described by the three Steinhart-Hart coefficients which are A, B, and C in Equation (1). However, as can be seen this curve is nonlinear which creates complications with both sensor circuit dynamic range and sensor calibration. If a temperature sensor is inherently linear or can be linearized, then a greater dynamic range over the entire operating temperature range can be realized, as is discussed in greater detail below.
                              1                      °K            .                          =                  A          +                      B            ⁢                                                  ⁢                          ln              ⁡                              (                R                )                                              +                      C            ⁢                                                  ⁢                                          ln                ⁡                                  (                  R                  )                                            3                                                          (        1        )            Equation (1) shows that a negative temperature coefficient of resistance (NTC) thermistor has an almost linear relationship between one over the temperature in degrees Kelvin (1/K) and the natural logarithm of the resistance, as is shown in FIG. 6.